Data-driven phase detector element for phase locked loops

ABSTRACT

Generating a composite interpolated phase-error signal for clock phase adjustment of a local oscillator by forming a summation of weighted phase-error signals generated using a matrix of partial phase comparators, each of which compare a phase of the local oscillator with a corresponding phase of a reference clock.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.17/327,512, filed May 21, 2021, naming Armin Tajalli, entitled “MatrixPhase Interpolator for Phase Locked Loop”, which is a continuation ofU.S. application Ser. No. 16/566,648, filed Sep. 10, 2019, naming ArminTajalli, entitled “Matrix Phase Interpolator for Phase Locked Loop”,which is a continuation of U.S. application Ser. No. 15/602,080, filedMay 22, 2017, naming Ali Hormati and Armin Tajalli, entitled“Data-Driven Phase Detector Element for Phase Locked Loop”, now U.S.Pat. No. 10,411,922, which claims the benefit of U.S. ProvisionalApplication No. 62/395,993, filed Sep. 16, 2016, entitled “Matrix PhaseDetector Element for Phase Locked Loop”, all of which are herebyincorporated herein by reference in their entirety for all purposes.

REFERENCES

The following prior applications are herein incorporated by reference intheir entirety for all purposes:

U.S. Patent Publication 2011/0268225 of application Ser. No. 12/784,414,filed May 20, 2010, naming Harm Cronie and Amin Shokrollahi, entitled“Orthogonal Differential Vector Signaling” (hereinafter “Cronie I”).

U.S. Patent Publication 2011/0302478 of application Ser. No. 12/982,777,filed Dec. 30, 2010, naming Harm Cronie and Amin Shokrollahi, entitled“Power and Pin Efficient Chip-to-Chip Communications with Common-ModeResilience and SSO Resilience” (hereinafter “Cronie II”).

U.S. patent application Ser. No. 13/030,027, filed Feb. 17, 2011, namingHarm Cronie, Amin Shokrollahi and Armin Tajalli, entitled “Methods andSystems for Noise Resilient, Pin-Efficient and Low Power Communicationswith Sparse Signaling Codes” (hereinafter “Cronie III”).

U.S. patent application Ser. No. 13/176,657, filed Jul. 5, 2011, namingHarm Cronie and Amin Shokrollahi, entitled “Methods and Systems forLow-power and Pin-efficient Communications with Superposition SignalingCodes” (hereinafter “Cronie IV”).

U.S. patent application Ser. No. 13/542,599, filed Jul. 5, 2012, namingArmin Tajalli, Harm Cronie, and Amin Shokrollahi entitled “Methods andCircuits for Efficient Processing and Detection of Balanced Codes”(hereafter called “Tajalli I”.)

U.S. patent application Ser. No. 13/842,740, filed Mar. 15, 2013, namingBrian Holden, Amin Shokrollahi and Anant Singh, entitled “Methods andSystems for Skew Tolerance in and Advanced Detectors for VectorSignaling Codes for Chip-to-Chip Communication”, hereinafter identifiedas [Holden I];

U.S. Provisional Patent Application No. 61/946,574, filed Feb. 28, 2014,naming Amin Shokrollahi, Brian Holden, and Richard Simpson, entitled“Clock Embedded Vector Signaling Codes”, hereinafter identified as[Shokrollahi I].

U.S. patent application Ser. No. 14/612,241, filed Aug. 4, 2015, namingAmin Shokrollahi, Ali Hormati, and Roger Ulrich, entitled “Method andApparatus for Low Power Chip-to-Chip Communications with Constrained ISIRatio”, hereinafter identified as [Shokrollahi II].

U.S. patent application Ser. No. 13/895,206, filed May 15, 2013, namingRoger Ulrich and Peter Hunt, entitled “Circuits for Efficient Detectionof Vector Signaling Codes for Chip-to-Chip Communications using Sums ofDifferences”, hereinafter identified as [Ulrich I].

U.S. patent application Ser. No. 14/816,896, filed Aug. 3, 2015, namingBrian Holden and Amin Shokrollahi, entitled “Orthogonal DifferentialVector Signaling Codes with Embedded Clock”, hereinafter identified as[Holden II].

U.S. patent application Ser. No. 14/926,958, filed Oct. 29, 2015, namingRichard Simpson, Andrew Stewart, and Ali Hormati, entitled “Clock DataAlignment System for Vector Signaling Code Communications Link”,hereinafter identified as [Stewart I].

U.S. patent application Ser. No. 14/925,686, filed Oct. 28, 2015, namingArmin Tajalli, entitled “Advanced Phase Interpolator”, hereinafteridentified as [Tajalli II].

U.S. Provisional Patent Application No. 62/286,717, filed Jan. 25, 2016,naming Armin Tajalli, entitled “Voltage Sampler Driver with EnhancedHigh-Frequency Gain”, hereinafter identified as [Tajalli III].

U.S. Provisional Patent Application No. 62/288,717, filed Apr. 22, 2016,naming Armin Tajalli, entitled “High Performance Phase Locked Loop”,hereinafter identified as [Tajalli IV].

The following additional references to prior art have been cited in thisapplication:

U.S. Pat. No. 6,509,773, filed Apr. 30, 2001 by Buchwald et al.,entitled “Phase interpolator device and method” (hereafter called[Buchwald].

“Linear phase detection using two-phase latch”, A. Tajalli, et al., IEEElectronic Letters, 2003, (hereafter called [Tajalli V].)

“A Low-Jitter Low-Phase-Noise 10-GHz Sub-Harmonically Injection-LockedPLL With Self-Aligned DLL in 65-nm CMOS Technology”, Hong-Yeh Chang,Yen-Liang Yeh, Yu-Cheng Liu, Meng-Han Li, and Kevin Chen, IEEETransactions on Microwave Theory and Techniques, Vol 62, No. 3, March2014 pp. 543-555, (hereafter called [Chang et al.])

“Low Phase Noise 77-GHz Fractional-N PLL with DLL-based ReferenceFrequency Multiplier for FMCW Radars”, Herman Jalli Ng, RainerStuhlberger, Linus Maurer, Thomas Sailer, and Andreas Stelzer,Proceedings of the 6th European Microwave Integrated CircuitsConference, 10-11 Oct. 2011, pp. 196-199, (hereafter called [Ng et al.])

“Design of Noise-Robust Clock and Data Recovery using anAdaptive-Bandwidth Mixed PLL/DLL”, Han-Yuan Tan, Doctoral Thesis,Harvard University November 2006, (hereafter called [Tan]).

U.S. Pat. No. 7,492,850, filed Aug. 31, 2005 by Christian Ivo Menolfiand Thomas Helmut Toifl, entitled “Phase locked loop apparatus withadjustable phase shift” (hereafter called [Menolfi].)

“A Calibration-Free Fractional-N Ring PLL Using HybridPhase/Current-Mode Phase Interpolation Method”, by Romesh Kumar Nandwanaet al, IEEE Journal of Solid-State Circuits Vol. 50, No. 4, April 2015,ppg. 882-895, (hereafter called [Nandwana].)

FIELD OF THE INVENTION

The present invention relates to communications systems circuitsgenerally, and more particularly to obtaining a stable, correctly phasedreceiver clock signal from a high-speed multi-wire interface used forchip-to-chip communication.

BACKGROUND

In modern digital systems, digital information has to be processed in areliable and efficient way. In this context, digital information is tobe understood as information available in discrete, i.e., discontinuousvalues. Bits, collection of bits, but also numbers from a finite set canbe used to represent digital information.

In most chip-to-chip, or device-to-device communication systems,communication takes place over a plurality of wires to increase theaggregate bandwidth. A single or pair of these wires may be referred toas a channel or link and multiple channels create a communication busbetween the electronic components. At the physical circuitry level, inchip-to-chip communication systems, buses are typically made ofelectrical conductors in the package between chips and motherboards, onprinted circuit boards (“PCBs”) boards or in cables and connectorsbetween PCBs. In high frequency applications, microstrip or striplinePCB traces may be used.

Common methods for transmitting signals over bus wires includesingle-ended and differential signaling methods. In applicationsrequiring high speed communications, those methods can be furtheroptimized in terms of power consumption and pin-efficiency, especiallyin high-speed communications. More recently, vector signaling methodshave been proposed to further optimize the trade-offs between powerconsumption, pin efficiency and noise robustness of chip-to-chipcommunication systems. In those vector signaling systems, digitalinformation at the transmitter is transformed into a differentrepresentation space in the form of a vector codeword that is chosen inorder to optimize the power consumption, pin-efficiency and speedtrade-offs based on the transmission channel properties andcommunication system design constraints. Herein, this process isreferred to as “encoding”. The encoded codeword is communicated as agroup of signals from the transmitter to one or more receivers. At areceiver, the received signals corresponding to the codeword aretransformed back into the original digital information representationspace. Herein, this process is referred to as “decoding”.

Regardless of the encoding method used, the received signals presentedto the receiving device must be sampled (or their signal value otherwiserecorded) at intervals best representing the original transmittedvalues, regardless of transmission channel delays, interference, andnoise. This Clock and Data Recovery (CDR) not only must determine theappropriate sample timing, but must continue to do so continuously,providing dynamic compensation for varying signal propagationconditions.

Many known CDR systems utilize a Phase-Locked Loop (PLL) or Delay-LockedLoop (DLL) to synthesize a local receive clock having an appropriatefrequency and phase for accurate receive data sampling.

BRIEF DESCRIPTION

To reliably detect the data values transmitted over a communicationssystem, a receiver must accurately measure the received signal valueamplitudes at carefully selected times. Various methods are known tofacilitate such receive measurements, including reception of one or morededicated clock signals associated with the transmitted data stream,extraction of clock signals embedded within the transmitted data stream,and synthesis of a local receive clock from known attributes of thecommunicated data stream.

In general, the receiver embodiments of such timing methods aredescribed as Clock-Data Recovery (CDR), often based on Phase-Lock Loop(PLL) or Delay-Locked Loop (DLL) synthesis of a local receive clockhaving the desired frequency and phase characteristics.

In both PLL and DLL embodiments, a Phase Detector compares the relativephase (and in some variations, the relative frequency) of a receivedreference signal and a local clock signal to produce an error signal,which is subsequently used to correct the phase and/or frequency of thelocal clock source and thus minimize the error. As this feedback loopbehavior will lead to a given PLL embodiment producing a fixed phaserelationship (as examples, 0 degrees or 90 degrees of phase offset)between the reference signal and the local clock, an additional fixed orvariable phase adjustment is often introduced to permit the phase offsetto be set to a different desired value (as one example, 45 degrees ofphase offset) to facilitate receiver data detection.

Methods and systems are described for receiving, at a data-driven phasecomparator circuit, a plurality of data signals in parallel from aplurality of multi-input comparators (MICs) connected to a multi-wirebus, wherein at least one MIC is connected to at least three wires ofthe multi-wire bus, and one or more phases of a local oscillator signal,the data-driven phase comparator circuit comprising a plurality ofpartial phase comparators, generating a plurality of partial phase-errorsignals using the partial phase comparators, each partial phase-errorsignal generated by receiving (i) a corresponding phase of the localoscillator signal and (ii) a corresponding data signal of the pluralityof data signals and responsive to a determination that a transitionoccurred in the corresponding data signal, generating the partialphase-error signal based on a comparison of the corresponding phase ofthe local oscillator signal and the corresponding data signal, andgenerating a composite phase-error signal by summing the plurality ofpartial phase error signals, the composite phase-error signal forsetting a local oscillator generating the one or more phases of thelocal oscillator signal in a lock condition.

Embodiments are described in which the Phase Detection and phaseadjustment elements are combined, leading to lower circuit nodecapacitance and reduced circuit delays, these improvements in turnenabling increased loop stability and improved PLL lock characteristics,including increased loop lock bandwidth leading to lower clock jitterand improved power supply noise rejection.

Embodiments are also described in which a Delay-Locked Loop is used toconvert the received reference clock signal into multiple referenceclock phases, converting the PLL phase comparison operation intomultiple comparisons made between a reference clock phase and a localclock phase. A summation or weighted summation of the multiplecomparison results is then used as the error feedback signal for thePLL. A further embodiment is described in which multiple comparisons aremade between a single received reference clock phase and multiple localclock phases, with the weighted sum of the multiple comparison resultsused as the error feedback term for the PLL. In at least one suchfurther embodiment, said weighted sums comprise a two dimensional timedomain filter.

BRIEF DESCRIPTION OF FIGURES

FIG. 1 is a block diagram of one embodiment capable of encoding andtransmitting five data bits and a clock on an eight wire communicationschannel.

FIG. 2 is a block diagram of one embodiment of a receiver compatiblewith the transmitter of FIG. 1.

FIG. 3 is a block diagram detailing one embodiment of the clock recoverycircuit used by the receiver of FIG. 2.

FIGS. 4A, 4B, and 4C show three Phase Detector embodiments suitable foruse in a Phase Locked Loop element of a clock recovery circuit.

FIG. 5 is a schematic diagram of one embodiment integrating an XOR phasedetector and clock phase interpolator.

FIG. 6A is a schematic diagram of a clocked data latch and FIG. 6B is aschematic diagram of a further embodiment of a clocked data latchintegrating a clock phase interpolator.

FIGS. 7A and 7B are schematic diagrams of an embodiment integrating astate machine phase detector and clock phase interpolators.

FIG. 8 is a schematic diagram of one embodiment of a charge pumpsuitable for further integration with a phase comparator embodiment.

FIG. 9 is a block diagram of a further embodiment in which multiplephases of a reference clock are compared with multiple local clockphases.

FIG. 10 is a block diagram of a further embodiment in which multiplecomparisons are made between a single reference clock and multiple localclock phases.

FIG. 11A is a weighted XOR phase detector, in accordance with someembodiments.

FIG. 11B is a block diagram of one embodiment of a matrix phasecomparison of M reference phases and N local clock phases.

FIGS. 12A and 12B are block diagrams of an alternate embodiment of theintegrated phase detector and phase interpolator of FIG. 5.

FIG. 13A is a timing diagram for a folded phase detector, in accordancewith some embodiments.

FIG. 13B is timing diagram illustrating a reverse clipping effect, inaccordance with some embodiments.

FIGS. 14A and 14B are timing diagrams for an array-XOR phase detectorand single-XOR phase detector, respectively, in accordance with someembodiments.

FIG. 15 illustrates XOR-based phase comparator and correction signalsapplied to a loop filter, in accordance with some embodiments.

FIG. 16 illustrates time-domain error signals produced by a row-basedphase comparator in accordance with some embodiments.

FIG. 17 illustrates phase interpolation based on interpolation a lockpoint between two consecutive diagonals of a two dimensional phasecomparator array, in accordance with some embodiments.

FIG. 18 illustrates improved resolution of a phase interpolator in adiagonal multi-phase detector structure, in accordance with someembodiments.

FIGS. 19A-19D illustrate various partial phase comparator architectures,in accordance with some embodiments.

FIG. 20 illustrates an exemplary XOR phase comparator architecture, inaccordance with some embodiments.

FIG. 21 illustrates a timing diagram of output currents Iout of an XORphase comparator shown in FIG. 20, in accordance with some embodiments.

FIG. 22 is a simulated phase comparator response, in accordance withsome embodiments.

FIG. 23 is a simulation of a phase-locked loop bandwidth, in accordancewith some embodiments.

FIG. 24 is a block diagram of an oversampled multi-phase feedbackphase-locked loop (MPLL) in accordance with some embodiments.

FIG. 25 is a block diagram of a receiver, in accordance with someembodiments.

FIG. 26 is a block diagram of a clock recovery circuit operating ondetected data signals, in accordance with some embodiments.

FIG. 27 is a block diagram of a matrix phase comparator operating ondetected data signals, in accordance with some embodiments.

FIG. 28 is a flowchart of a method, in accordance with some embodiments.

FIG. 29 is a block diagram of an edge-triggered bang-bang phasedetector.

FIG. 30 is a block diagram of a linear edge-triggered phase detector.

FIG. 31 is a wave-form associated with the linear edge-triggered phasedetector of FIG. 30.

DETAILED DESCRIPTION

As described in [Cronie I], [Cronie II], [Cronie III] and [Cronie IV],vector signaling codes may be used to produce extremely high bandwidthdata communications links, such as between two integrated circuitdevices in a system. As illustrated by the embodiment of FIG. 1,multiple data communications channels transmit symbols of the vectorsignaling code, acting together to communicate codewords of the vectorsignaling code. Depending on the particular vector signaling code used,the number of channels comprising a communications link may range fromtwo to eight or more, and may also communicate one or more clock signalson separate communications channels or as sub-channel components of thevector signaling code. In the example of FIG. 1, communication link 120is illustrated as being composed of eight wires 125, collectivelycommunicating five data values 100 and one clock 105 between transmitter110 and receiver 130.

Individual symbols, e.g. transmissions on any single communicationschannel, may utilize multiple signal levels, often three or more.Operation at channel rates exceeding 10 Gbps may further complicatereceive behavior by requiring deeply pipelined or parallelized signalprocessing, precluding reception methods that include the previousreceived value to be known as the current value is being received.

Embodiments described herein can also be applied to prior artpermutation sorting methods not covered by the vector processing methodsof [Cronie II], [Cronie III], [Cronie IV], and/or [Tajalli I]. Moregenerally, embodiments may apply to any communication or storage methodsrequiring coordination of multiple channels or elements of the channelto produce a coherent aggregate result.

Receiver Data Detection

To provide context for the following examples, one typical high-speedreceiver embodiment [Stewart I] is used for illustrative purposes,without limitation.

As illustrated in FIG. 2, the example data receiver includes eightidentical Continuous Time Linear Equalization (CTLE) stages 210operating on the signals received on the eight wires, previously shownas 120 in FIG. 1.

As described in [Tajalli I], [Holden I] and [Ulrich I], vector signalingcodes may be efficiently detected by linearly combining sets of inputsignals using Multi-Input comparators or mixers (MIC). For the 5b6w codeused by the example receiver, five such mixers acting on weightedsubsets of the six received data input signals will detect the five databits without need of further decoding. One additional mixer acting oncombinations of the two received clock signals will similarly detect theclock signal. In FIG. 2, this set of six MIC mixers 220 operate on thereceived and equalized signals to produce detected signals MIC0-MIC5.

Because of the high data rates involved, multiple parallel phases ofreceive processing are shown in the example receiver. In one embodiment,the five detected data signals MIC0-MIC4 are processed in four parallelphases of receive data processing, each phase 230 including five datasamplers and subsequent buffering, followed by recombination of the fourphase outputs into a received data stream, shown in FIG. 2 as beingperformed by multiplexers 240.

Clock Recovery circuits (also known in the art as Clock Data Recovery orCDR) support such sampling measurements by extracting timinginformation, either from the data lines themselves or from dedicatedclock signal inputs, and utilize that extracted information to generateclock signals to control the time interval used by the data linesampling device(s). The actual clock extraction may be performed usingwell known circuits such as a Phase Locked Loop (PLL) or Delay LockedLoop (DLL), which in their operation may also generate higher frequencyinternal clocks, multiple clock phases, etc. in support of receiveroperation. In the embodiment of FIG. 2, the detected clock signal isobtained at MIC5 and processed 300 to extract properly timed samplingclocks for the four data phases.

Other embodiments may forgo the dedicated wires used to communicate aseparate clock signal, and instead be configured to have the receiverextract a clock from transitions occurring on the data lines themselves.As is well understood in the art, successful application of thistechnique requires a sufficiently large transition density on the datalines (which may be alternatively interpreted as requiring asufficiently small interval between transitions,) and sufficientfree-running frequency stability within the PLL to maintain accuratedata sample timing during non-transition intervals. [Shokrollahi I]describes suitable vector signaling codes having such guaranteedtransition density. Alternatively, known art transition-enforcingencoding such as the commonly utilized 8b10b and 64b66b codes may beapplied to the data stream (or data streams of individual subchannels)to ensure subsequent transitions in the vector signal code elements oneach subchannel to insure a guaranteed minimum transition density at thereceiver.

PLL Overview

Phase Locked Loops are well represented in the literature. A typical PLLis composed of a phase detector that compares an external referencesignal to an internal clock signal, a low pass filter that smooths theresulting error value to produce a clock control signal, and a variablefrequency clock source (typically, a Voltage Controlled Oscillator orVCO) controlled by the smoothed error value, producing the internalclock signal presented to the phase detector. In a well-know variation,such a PLL design may incorporate a clock frequency divider between theVCO and the phase detector, allowing a higher-frequency clock output tobe phase locked to a lower-frequency reference signal.

In an alternative embodiment, the variable frequency clock source isreplaced by a variable delay element, its (optionally multiple tapped)outputs thus representing one or more successive time-delayed versionsof the original input signal rather than successive cycles of anoscillator to be phase compared to the reference input signal. For thepurposes of this document, such Delay Locked Loops (DLL) are consideredfunctionally equivalent to a PLL in such an application, particularly inregard to composed elements of phase detector, phase interpolator, andcharge pump.

Numerous forms of phase detectors are known to the art. A simple XORgate as in FIG. 4A may be used to compare, as a non-limiting example,two square wave signals. One familiar with the art will observe thatsuch a digital XOR output will be a variable-duty-cycle waveform which,when low pass filtered into an analog error signal, results in aproportional error signal centered in its analog signal range when thetwo input signals have a 90 degree phase offset relationship.

The more complex state machine phase detector of FIG. 4B is composed oftwo edge-triggered latches clocked respectively by the reference andinternal clock signals, with the first received clock edge initiating anoutput signal on one of the “early” or “late” outputs. Either outputbecoming active will subsequently cause the latches to reset inanticipation of the next comparison interval. Alternative embodimentsmay incorporate a timing delay in this reset path to provide additionalcontrol of the reset pulse timing, as indicated by the “Hold” signal.The “late” and “early” phase comparison outputs are typically acceptedas “pump up” and “pump down” inputs to a charge pump, the output ofwhich is the analog error value. That is, a pump up signal may turn on afirst transistor circuit that provides charge to capacitor therebyincreasing the analog voltage, while a pump down signal may turn on asecond transistor circuit that removes charge from a capacitor, therebyreducing the voltage. A zero degree phase offset between the two inputclock signals will thus leave the analog error value unchanged and thePLL in a stable locked condition. A number of equivalent state machinephase detector embodiments are known in the art, and are equallyapplicable in this application, without implication of limitation. Somestate machine embodiments may be sensitive to both phase and frequencydifferences between the input signals, facilitating more rapid PLL lockacquisition on startup.

As shown in FIG. 4C, a simple edge-clocked “D” flip-flop may also beused as a phase detector. At each rising local clock edge (CkPLL), the Dinput samples the state of the (in this example, square wave) referenceinput (CkRef); if it is high (e.g. it has already transitioned,) the Qoutput is high indicating the reference is “early”, if it is low (e.g.it has not yet transitioned,) Q is low indicating the reference is“late”. This so-called “bang/bang” phase detector provides a lessnuanced error result than the previous example, thus may utilize moresophisticated filtering to obtain loop stability.

As will be recognized by those familiar with the art, comparablefunctional operation may be obtained regardless of the phase detectortype incorporated in a PLL design, thus to first approximation phasedetector choice is not limiting. Secondary design behaviors, includinglock time, stability, power consumption, etc. must also be considered aspart of the design process.

Receiver Clock Recovery

The example receiver utilizes a PLL embodiment as shown in FIG. 3. ThisPLL accepts the received clock signal R5 as the reference to which itsclocks will be phased locked. In some embodiments, logic level shift 310is used if appropriate to interface between the signal levels providedby the detecting MIC and the preferred phase comparator input levels.Phase Comparator 320 compares the reference clock to an internal clockderived from the VCO, producing an output which is low pass filtered toprovide an Error value which subsequently corrects the operatingfrequency of VCO 340. In some embodiments, the outputs of PhaseComparator 320 is a digital waveform requiring conversion to an analogerror signal, either through implicit or explicit digital to analogconversion, or by use of an interface element such as a charge pump.Some embodiments may combine such conversion with all or part of the lowpass filtering operation, as one example offered without limitation, bythe digital filtering behavior shown by the switching action of a chargepump directed by digital control signals generating an analog signaloutput.

In one embodiment, a ring oscillator 340 composed of a sequence ofidentical gates in a closed loop is used as the internal VoltageControlled Oscillator (VCO) timing source for the PLL. The VCO frequencyis varied by analog adjustment of at least one of: gate propagationdelay, inter-gate rise and fall time, and gate switching thresholdwithin the ring oscillator. This may be implemented via switchedcapacitor banks, where a digital control signal is applied to selectiveplace capacitive elements in parallel and/or series combinations toalter an RC time constant, as one non-limiting example. Still further, acurrent source that drives a gate of the ring oscillator may beincreased or decreased to alter the output switchingrise-time/fall-time, and thereby adjust the effective delay. Outputstaken at equal intervals (i.e. separated by equal numbers of ringoscillator gates) along the sequence of gates comprising the ringoscillator provide the four data phase sampling clocks, hereinidentified as the 0, 90, 180, and 270 degree clocks.

In one embodiment, the ring oscillator is composed of eight identicalsets of logic gates (e.g., a set of inverter circuits), thus the phasedifference from one such set to the next is 45 degrees. In thisembodiment, the 0, 90, 180, and 270 degree outputs may be obtained, asexamples, from the second, fourth, sixth, and eighth outputs. As theseclocks are cyclical, the final tap may be considered as logicallyadjacent to the initial tap, a 0 degree and a 360 degree phase offsetbeing equivalent. As many variations of such designs are known in theart, neither the number of elements in the ring oscillator nor thespecific taps at which particular outputs are made should be construedas implying a limitation. As one example, the location of the 0 degreetap is arbitrary, as one familiar with the art will recognize thatnormal PLL behavior will phase align the ring oscillator with theexternal phase reference regardless of its initial phase. Similarly,equivalent designs may be obtained in which the output clock phases donot have square wave duty cycles; as one example being produced by theaction of AND or OR gates with inputs from different tap locations. Inthe example receiver, it is desired that the VCO operate at a multipleof the received reference clock frequency, thus Frequency Divider 350divides the VCO outputs by a comparable amount prior to the PhaseDetector. In one embodiment, binary (factor of two) dividers are used at350 to obtain the correct sampling clock rate. In another embodiment, nodivider is utilized and the VCO outputs are presented to the phaseinterpolator directly.

Each of the four phases of sampling clocks is appropriately timed tosample received data for one of the four parallel processing phases. Inparticular, internal clock ph000 is aligned to optimally trigger datasamplers in the phase0 phase of processing, clock ph090 in phase1, clockph180 in phase2, and clock ph270 in phase3.

To allow the overall phase of the locked PLL signals to be offset fromthe reference clock input phase, the local clock output presented to thephase comparator is obtained from phase interpolator 360, the outputphase of which is controllably intermediate between its input clockphases. Thus, the PLL may lock with its fixed phase relationship, whilethe internal clock signals obtained from ring oscillator 340 will beoffset from that fixed phase by the phase delay amount introduced byphase interpolator 350, as controlled by signal Phase offset correctiongenerated by clock/data phase control logic 370. Phase interpolators areknown in the art, examples being provided by [Buchwald I] and [TajalliII].

In one embodiment, phase interpolator 360 receives multiple clock phasesfrom the ring oscillator 340 having 90 degree phase differences. Saidphase interpolator may be controlled to select two adjacent clock inputphases and then to interpolate between them so as to produce an outputat a chosen phase offset between those selected two values. For purposesof description, it may be assumed that a phase detector design is usedwhich drives the PLL to lock with a zero phase differential between thetwo phase detector inputs. Thus, continuing the example, applying the 0and 90 degree clock phases as inputs to the phase interpolator allowsadjustment such that the PLL leads the reference clock input by between0 and 90 degrees.

It will be apparent that equivalent results with comparable phaseoffsets may be obtained using other pairs of degree clocks and/or otherphase detector designs, which as previously described may lock withdifferent phase differentials than that of the present example. Thusneither the particular phase clocks chosen nor the particular phasedetector design described herein are limiting.

In the known art, [Nandwana] describes a Fractional-N clock multiplyingPLL in which a single reference clock is phase compared to two localclocks derived using different integer divisor ratios, withinterpolation between the two phase error results dynamically chosen tocancel the phase quantization error.

Phase Detector with Interpolator

As communication channel data rates increase, it becomes increasinglydifficult to maintain acceptable PLL lock range and accuracy, asinherent and parasitic circuit node capacitances introduce circuitdelays and constrain the effective loop response bandwidth. Anembodiment providing improved response characteristics suitable for suchhigh speed operation is illustrated in FIG. 5. As one familiar with theart will observe, this is a CMOS design providing symmetrical operationfor both positive and negative output excursions, integrating elementsof both phase interpolator and phase detector designs. This tightintegration results in reduced node capacitances, facilitating thedesirable high speed operation, and the balanced differential structuresimplifies the control of charge and discharge currents.

As with conventional designs, the PLL VCO (or a clock divider driven bysaid VCO) provides the local oscillator inputs to phase interpolatorelements 510 and 515, which together set the effective local clockphase. Four local oscillator phases with 90 degree offset are shown i.e.equivalent to two phases in quadrature relationship and theircomplimentary signals and thus identified as +I, +Q, and −I, −Q,permitting a full 360 degree or “four quadrant” phase adjustment. Otherembodiments may utilize as few as two local oscillator phases, may useoscillator phases having other than 90 degree phase differences, or mayselect clock phases from an input set of more than four; as onenon-limiting example, choosing at least two clock phases to beinterpolated between from an input set of eight clock phases.

In a first embodiment, phase interpolator element 510 includes fourmixing elements, each mixing element comprising a differentialtransistor pair and a controlled current source, with a commondifferential output driven by the four mixing elements in parallel.Thus, configuration of current source IA(i) controls the amount of localoscillator phase +I presented to the common output ckp; similarly,current source IA(−i) controls the amount of complimentary output phase−I in the output, IA(q) controls the amount of +Q, and IA(−q) controlsthe amount of −Q. It will be readily apparent to one familiar with theart that configuration of the four current sources can produce an outputclock at Ckp having any desired phase relationship to the PLL localclock input.

Similarly, phase interpolator element 515 current sources IB(i), IB(−i),IB(q), and IB(−q) may be configured to obtain an output clock at Cknhaving any desired phase relationship to the PLL local clock input.Typically, CkPLLp and CkPLLn will be configured to have complimentaryrelationships so as to provide phase detector 520 with balanced andcomplimentary positive- and negative-going current amplitudes. However,configuration with non-complimentary IA and D3 values may be performedto obtain particular results. As one example offered without limitation,an embodiment separately adjusting IA and IB values might obtain higherresolution phase adjustment, compared to an embodiment maintainingperfectly complimentary IA and IB values.

The second input to the Phase Detector 520 is external reference clockCkRef+/CkRef−, producing the phase error output currentsVCOctl+/VCOctl−. In one advanced embodiment, the two external referenceclocks are of opposing polarity but not necessarily complementary phase,thus the positive polarity comparison and negative polarity comparisonrepresent different phase comparisons. Such an advanced embodiment maybe combined with non-complimentary IA and IB bias configurations,providing independent adjustment of local clock phase during thosedifferent phase comparisons. That is, in one embodiment, the CkRef inputat the top of PD 520 is a first phase selected from the reference clockphases available in the circuit, and the IA currents are adjusted toprovide a corresponding interpolated phase offset from the firstselected phase, and the CkRef input at the bottom of PD 520 is a secondphase selected from the reference clock phases available in the circuit,and the IB currents are adjusted to provide a corresponding interpolatedphase offset from the second selected phase, wherein the amount of therelative phase offsets are the same.

Configuration of phase interpolator current source values may beperformed by external control logic, including without limitation, ahardware configuration register, control processor output register, andhardware CDR adjustment logic.

Alternative Phase Detector Embodiments

Phase Detector 520 in the embodiment of FIG. 5 is shown as an XOR-styledevice as in FIG. 4A, mixing local clock CkPLL and external referenceclock CkRef to produce phase error output VCOctl. In the alternativeembodiment of FIG. 12A, a folded phase detector is used at 1220, drivenby currents produced by the combination of phase interpolator 510 andcurrent sink Ifix2, and the combination of phase interpolator 520 andcurrent source Ifix1. The folded phase detector embodiment shown in FIG.12A is described in further detail below. As with thepreviously-described embodiment, current sources IA(i), IA(−i), IA(q),and IA(−q) are configured to produce the desired interpolation of PLLclocks i, q, and −q in interpolator outputs CkPLLp and CkPLLp, whilecurrent sources IB(i), IB(−i), IB(q), and IB(−q) are configured toproduce the desired interpolation of PLL clocks i, q, and −q ininterpolator outputs CkPLLn and CkPLLn. Phase comparator 1220 is alsodriven by received reference clocks CkRef+ and CkRef−, producing phasecomparison results Phase Error (+) and Phase Error (−). In someembodiments, the circuit node labeled Circuit Balance Feedback may bemonitored to determine the relative DC component of the interpolatedclock signals, which may then be modified by adjustment of theconfigured current source values in 510 and 515. In some embodiments,each current source IA and IB receives seven control bits. It should benoted that embodiments are not limited to receiving seven control bits,and that any number of control bits may be implemented according todesign constraints for PI resolution, for example. In some embodiments,current sources IA and IB are equal (e.g., IA=IB for +/−q). In suchembodiments, the PIs 510 and 515 have 7 bits of resolution. Inalternative embodiments, additional resolution may be implemented byintroducing a shift in IB with respect to IA, or vice versa. In anexemplary embodiment, IA=IB+8, where 8 is a decimal shift added to thecontrol bits of each current source IA to obtain the control bits ofeach current source IB. In such embodiments, the P-side PI 510 andN-side PI 515 are looking into two different VCO phases, and the phasedetector collects information from different phases of the VCO. Sincethe PIs 510 and 515 combine information from different phases of VCO,the PLL has more detailed information about phases of PLL and thebandwidth of the PLL is higher than a conventional PLL.

Embodiments for which IA=IB+shift are a special case of a matrix phasecomparator in which there are two phase comparators. The first phasecomparator (NMOS-side XOR) compares the phase of reference with one setof VCO feedback phases, and a second phase comparator (PMOS-side XOR)that compares the reference clock phase with a second set of VCOfeedback phases. Unlike the [Nandwana] phase comparator, the sets of VCOfeedback phases here are of the same frequency, differing only in phase,and the current source values chosen to interpolate between the phasecomparison results will typically be static rather than dynamicallychosen on a cycle-by-cycle basis. Matrix phase comparator embodimentsare described in further detail below. Therefore, in some embodiments, aPMOS+NMOS interpolator may be treated as two independent PIs, while in[Nandwana], there is only one PI. Further, if a meaningful difference tothe weighs of PMOS-side and NMOS-side, then a small matrix PLL may beconstructed that has extended bandwidth. In at least one embodiments,there is a 20% gain difference between the two sides, in which BW mayenhance by the same factor.

In some embodiments, a folded structure as shown in FIG. 12A may beused. FIG. 12A is similar to the embodiment shown in FIG. 5, however thephase detector 520 is replaced with a folded phase detector 1220. Asshown, folded phase detector 1220 includes current sources Ifix1 andIfix2, which may be configured to provide more voltage headroom to thePMOS PI current sources IA and the NMOS PI current sources IB. Further,phase detector 1220 includes a pair of transistor branches connected toCkPLLp and CkPLLn. For purposes of illustration, consider PI 510 and 515only having IA(i) and IB(i) turned on respectively, representing phaseph0000 from the VCO. In the case where CkRef is offset 90 degrees fromph0000, the folded phase detector 1220 will be in lock condition. Asshown in FIG. 13A, during the first 180 degrees (1) of a period, for afirst 90 degrees (2), current Ip is charged to the (−) terminal of thePhase Error signal through transistor 1206 using PMOS PI 510. At thesame time, current In is discharged from the (−) terminal of the PhaseError signal through transistor 1208 using NMOS PI 515. Similarly,during the second 90 degrees (3), current Ip is charged from the (+)terminal of the Phase error signal through transistor 1202, whilecurrent In is discharged from the (+) terminal through transistor 1204.As shown, Ifix2 will sink a fixed amount of current being provided fromPMOS PI 510, while Ifix 1 sources some current to NMOS PI 515 to preventthe current sources in the NMOS PI from sinking too much current fromthe Phase Error signal. Such a technique provides a reverse clippingeffect. One of skill in the art may notice that equally adjusting theIfix current magnitudes may have an effect on the range of the PhaseError signal. In some embodiments, increasing the Ifix magnitudes willlower the magnitude range of the Phase Error signal, while decreasingthe Ifix magnitudes will increase the magnitude range of the Phase Errorsignal. This relationship can be found in FIG. 13B.

FIG. 13B is a timing diagram illustrating the reverse clipping featuredescribed above. FIG. 13B depicts the magnitude of current Ip in thefirst 180 degrees (1) for two values of Ifix2: A and B, where A>B. Asshown, the magnitude of Ip is less in the case of Ifix2=A. When Ifix2=B,the magnitude range of Ip is relatively higher. One of skill in the artwould notice a similar effect occurs in the case of In being dischargedfrom the folded phase detector 1220.

In some embodiments, the second 180 degrees (4) may be used to providecircuit balance feedback, as shown in FIG. 12A. During the circuitbalance feedback phase (4), current may be charged via the PMOS PI 510while current is discharged via the NMOS PI 515. If there is animbalance of charge/discharge currents, a non-zero circuit balancefeedback signal may indicate this imbalance, which may occur due totransistor mismatches, for example. The circuit balance feedback signalmay then be used to adjust either Ifix1 or Ifix2 to balance thecharge/discharge currents so that the balance feedback signal is zero.In some embodiments, the voltages of the charge-pump circuit may bemonitored, and if equal, the circuit is properly balanced, i.e., Ip=In.A simplified schematic of the phase comparator circuit of FIG. 12A isshown in FIG. 12B.

The phase detector of [Tajalli V] may alternatively be used at 520 or1220, providing equivalent phase detection with enhanced signal headroomin embodiments utilizing low power supply voltages. Other phasedetectors, including all variations shown in FIGS. 4A, 4B, and 4C, mayalso be substituted at 520 in that embodiment.

As one example of such alternative embodiment, the State MachinePhase/Frequency Detector of FIG. 4B may be combined with the PhaseInterpolator design of FIG. 5.

FIG. 6A shows a schematic of one embodiment of a conventional CIVILclocked latch, composed of a clocked feedback latch outputting results Qand Q the state of which is initialized by clocked differential inputs Dand D. FIG. 6B shows the same circuit in which the clock source phase ismodified by phase interpolator 615, operation of which is as previouslydescribed for FIG. 5.

Substituting the clocked latch circuit of FIG. 6B into each D flip-flopinstance of FIG. 4B produces the alternative embodiment shown in FIGS.7A and 7B. D flip-flop 710 is clocked by the received clock CkRef, whichis passed through phase interpolator 715. As an example and for purposesof explanation, without a configured phase offset (or a desired offsetof 0 degrees), current source IA would be set to “mix” input CkRef at100% proportion, and the other three current sources set to zerocurrent. D flip-flop 720 is clocked by local clock CkPLL, which isobtained by configuration of phase interpolator 725 current sourcesIB(i), IB(−i), IB(q), and IB(−q), which in turn controls the relativeproportions and polarities of I and Q clocks being combined. In oneembodiment, I is obtained from ph000, −I from ph180, Q from ph090, and−Q from ph270, as seen in FIG. 3. A simple CIVIL OR gate 730 drives thereset function for flip-flops 710 and 720.

It should be noted that in this one embodiment the majority of phaseinterpolator 715 is functionally disabled and retained only to preservethe same parasitic load characteristics as are presented by active phaseinterpolator 725, to maximize circuit symmetry and maintain balancedloading characteristics to minimize secondary effects such as detectionbias and drift.

Integrated Phase Detector, Interpolation, and Charge Pump

As previously described, PLL phase detector outputs are typically usedto drive a charge pump circuit, the output of which is an analog errorsignal used to control the VCO. The described improvement from reducedcapacitance and resulting higher circuit speed in integrating the PLLphase detector and clock adjustment phase interpolator may be furtherextended by also integrating elements of the charge pump in the samemanner.

In this combined embodiment, the charge pump control signals UPp, UPn,DOWNp, and DOWNn provided by the embodiment shown in FIGS. 7A and 7Bdirectly control the charge pump embodiment of FIG. 8 to produce outputIOUT. Current source ICPC and voltage reference VREF may be configuredto scale and adjust the IOUT range. One familiar with the art will notethe significant symmetry in the circuit of FIG. 8, allowing accuratetracking between generation of VREPLICA and IOUT signals.

FIG. 8 is a schematic of a charge pump circuit with improvedcharge/discharge current balancing, in accordance with some embodiments.The circuit 800 includes two parallel charge pumps 802, 804: the twodifferential pairs within charge pump 804 generate an output currentrepresenting a phase error in response to the up and down pulses, andthe two differential pairs of charge pump 802 are used to set thedischarge current to be equal to the charge current as described below.Specifically, the current source ICPC sets a charging current level byproviding a corresponding bias voltage VBP through a current mirroringcircuit to drive the top current sources 806, 808 of the two chargepumps so as to also provide ICPC to each charge pump 802, 804. When UPngoes low and turns on FET 810, the node 812 is charged (capacitiveelement 814 is either a discrete cap or a parasitic cap) by the chargingcurrent ICPC provided by FETs 806, 810. In a balanced condition (i.e.,in the absence of a phase error), the amount of current that is thendischarged during a high DOWNp signal through the bottom FET 816 shouldbring the node 812 back to the VREF value. If the discharge current istoo low and the voltage VREPLICA increases above VREF, then theamplifier 820 will increase the bias voltage VBN to thedischarge-current FET 818 to increase the amount of discharge current sothat it equals the charge current ICPC and the voltage VREPLICA at node812 is brought back to VREF. On the other hand, if the discharge currentset by VBN on FET 818 is too high, the VREPLICA voltage drops too low,and the amplifier 820 responsively reduces the bias voltage VBN ondischarge-FET 818 to bring the charge pump currents into equilibrium.

Second order PLLs (called also charge pump PLLs) have been widely usedto implement low noise and high performance synthesizers, clockgenerators, and clock and data recovery systems. A phase detector (PD),or a phase-frequency detector (PFD) produces a signal proportional tothe phase difference between the reference clock (CkRef) and thefeedback clock (CkPLL). The resulting error is integrated by charge-pumpcircuit (CPC) and loop filter (LF) to produce the proper control voltagefor the voltage (or sometimes current) controlled oscillator (VCO). Anexemplary LF is an RC circuit as shown in FIG. 24. Many modernintegrated VCOs are based on differential topology that can provide twocomplementary outputs. A differential architecture provides moreresistivity against supply and substrate noise. LC tank based VCOs andring oscillators are two main categories of controlled oscillators thathave been very widely used in high speed communication systems. Bothtopologies can be configured to provide two or more output phases, whichis essential for multiphase systems and facilitates the enhancementsdescribed below.

Other embodiments may be obtained by equivalent combination of phasecomparator, phase interpolator, and charge pump elements.

Oversampling of Input Reference Signal

The asymmetric use of the phase interpolators in, as one example, FIGS.7A and 7B, stems from the different nature of the local clock andreference clock sources. The former is obtained from a multiphase clocksource (e.g. an oscillator or divider) inherently capable of providingthe multiphase inputs utilized by a phase interpolation element. Thelatter is generally single phased, obtained from (typically) onereceived clock source.

In the known art, [Tan] described a combined DLL/PLL structure, in whichthe voltage controlled delay line incorporated in the PLL VCO isduplicated as an input delay line acting on the reference clock input,and controlled by a single feedback error signal. [Ng] and [Chang] alsodescribe use of a front-end DLL to serve as a frequency multiplier tofacilitate generation of very high frequency clocks.

However, if such a controlled delay line is tapped, and so configuredthat the differential delay between taps is proportional to the timebetween received clock edges, a received clock passing through such adelay line produces a resulting set of outputs which take on some of thecharacteristics of a multiphase clock. As one example offered withoutlimitation, the equal-interval outputs of a four tap delay line havingan overall delay comparable to the reference clock period will provideoutputs having similar characteristic to quadrature phased clocksignals. Continuing this example, if each such output is phase comparedto an appropriately-selected local clock phase, a series of phase errorresults will be produced which may be combined to produce a moreaccurate aggregate clock error signal for the PLL VCO. The delayedversions of the receive clock represent additional opportunities forphase comparison with a clock derived from the VCO, thus providing ahigher update rate for the controlled loop, and thus improved PLL loopbandwidth leading to reduced jitter and better noise immunity. That is,using this technique, the update rate of the loop will be increased,which in turn enables the circuit to track and correct the effects ofnoise and jitter at higher frequencies.

For the delayed phase comparisons to provide meaningful information tothe PLL, the delay intervals provided by the delay line must becoordinated with the period between local clock phases, with suchcontrols giving the delay element many of the aspects of a Delay-LockedLoop (DLL.) As seen in the block diagram of FIG. 9, the external clockreference input to the previous PLL embodiment 300 is provided by DLL910. The received clock signal R5 is presented to tapped delay line 916,producing a series of received clock phases 918. The DLL control loop isprovided by phase comparator 912 comparing the received clock with adelayed clock, producing an error value that is Low Pass Filtered 915,producing a Delay Adjust signal controlling the delay line timing.

Within PLL 300, the previous simple phase comparison (320 of FIG. 3) isnow performed by multi-phase comparison 920. In one embodiment, XORgates compare the phase of each received reference clock phase on the Nlines (e.g., N=2, 4, 8, etc., and possibly including odd integers aswell to obtain other phases such as 60, 120, 180, 240, 300) 918 with adifferent clock phase from the N lines 965 from phase interpolator 360.Each XOR gate output is converted to an analog signal value, all suchanalog signal values being summed to produce a composite analog Errorresult controlling ring oscillator 340, as previously described. In afurther embodiment, summation 935 is performed by a weighted summationnode comparable to the previously-described MIC mixer, the differentselected weights of said summation allowing further control of PLLstatic and dynamic operational characteristics. Alternatively, each XORoutput can be used to drive a separate transistor circuit for injectingor removing charge from a capacitive element to achieve the summation.In a further embodiment, each XOR phase comparator may include aplurality of AND operations implemented as transistor branches, each ANDoperation configured to provide a current output to a common summationnode, the magnitude of each current being independently configurable soas to provide a weighting function to each AND operation. In addition,the PLL 300 of FIG. 9 may be configured to provide a desired phaseoffset, where the interpolated phases each have the same offset relativethe tap delay line signal to which it will be XOR compared.

In some system environments, the described multi-phase reference clockmay be directly available from the receiver, as one example where thecommunications protocol incorporates multiple clock signals.

The additional feedback information provided by the multiple comparisonoperations may also be obtained without the previously-described DLLfront end. FIG. 10 shows an embodiment in which the single receivedreference signal 1018 enters multi-phase comparator 920 in which thesingle received reference signal is compared to each of two or morelocal clock phases 965. As in the previous example, this multiple phasecomparison is distinct from that of [Nandwana] in that all of the localclock phases used for comparison are of the same frequency, differingonly in phase. In one embodiment, XOR gates compare the phase of thesingle received reference clock phase 918 with a different clock phase965 from phase interpolator 360. Each XOR gate output is converted to ananalog signal value, all such analog signal values being summed toproduce a composite analog Error result controlling ring oscillator 340,as previously described. In a further embodiment, summation 935 isperformed by a weighted summation node comparable to the previouslydescribed MIC mixer, the different selected weights of said summationallowing further control of PLL static and dynamic operationalcharacteristics. In another embodiment, each XOR phase comparatorprovides a current output to a common summation node, the magnitude ofeach current being configurable so as to provide a weighting function.In particular, such weight adjustments may be used to produce additionalclosed-loop poles and/or zeroes in the PLL time domain transferfunction, providing additional control of loop stability.

FIG. 14A is a timing diagram of a reference signal CKREF being comparedwith four phases of the VCO (feedback from the PLL):

XOR(CKREF, VCO'000) XOR(CKREF, VCO'045) XOR(CKREF, VCO'090) XOR(CKREF,VCO'135)

As shown in FIG. 14A, it is assumed all weights are equal, however thisis purely for illustrative purposes, and should not be consideredlimiting in any way. FIG. 14A further includes a summation of the fourXOR outputs. As can be seen, in lock condition, the integral of thebottom waveform is zero, and the PLL will lock properly. Forconvenience, FIG. 14B has been included to illustrate a conventional XORbased phase detectors in which the reference is compared to only one VCOphase. In lock position, the reference and VCO are 90-degree phaseshifted, and the output of XOR is a rectangular waveform with an averagevalue equal to zero. One may observe how the two waveforms (simple XORin the FIG. 14B and array-XOR in FIG. 14A) differ from each other;however in both cases the average value for a given period is zero, andthe PLL locks. In embodiments utilizing an array phase detector, alarger number of transitions occur with respect to a single XOR phasedetector. As each transition carries information about an edge, a largernumber of transitions means that phase comparator has been able tocollect more information from VCO and CKREF.

It should be noted that in array-XOR embodiments, some comparisons mightbe done using XNORs. As such, an XOR or XNOR for different phasecomparisons may be selected carefully to ensure system stability.

In at least one embodiment, the weights of said summation are configuredsuch that they decline in proportion to the timing difference of thecomparison clock phase relative to the PLL “normal lock” phase. As oneexample offered without limitation, if ph090 is the normal lock phase ofthe PLL, the comparison of ph090 and the received reference signal isweighted 1; comparisons of ph045 and ph135 (e.g. a half tap offset fromthe normal lock phase) are weighted ½; comparison of the receivedreference signal and ph000 and 180 (one tap offset from the normal lockphase) are weighed ¼; etc. These various weighted comparison results arethen summed to produce a composite signal which when low pass filtered330, is the Error value controlling PLL VCO 340.

In one embodiment utilizing multiple phase comparators, thedeterministic jitter produced by the multiple phase comparisons was seento occur at a 12.5 GHz rate with equal phase detector weights. Eventhough the amount of jitter was very small and the jitter rate was wellabove the loop filter cutoff frequency, the deterministic jitter wassignificantly reduced with the described weight adjustments, in whichweight magnitudes decline in proportion to their offset distance fromthe primary reference signal sample. In some embodiments, differentweighted values are used in a comparator circuit to construct a discretetime domain filter. This property can be used to simplify the design ofanalog filter 330. For example, with proper weighting values one mightconstruct a discrete time domain zero in the transfer function thatprovides conditions to make the loop robust.

As with previously described examples, other embodiments may be obtainedby equivalent combination of phase comparator, phase interpolator, andcharge pump elements.

Matrix Phase Comparisons

In some embodiments, BW of a PLL is limited by the update rate of theloop, which is determined by the frequency of the reference clock.Certainly, using all the available sources of information in system cansubstantially enhance the efficiency of the correction loop. Forexample, every phase of the VCO provides a single sample of theoscillator phase during each cycle of the reference clock period, whilelooking into all the phases of the VCO can provide more detailedinformation in the time span of Tref shown in FIG. 14B. In conventionalPLLs, only one of the VCO phases is fed back into the phase detector.Hence, the phase detector has only part of the available informationregarding instantaneous phase of oscillator. The following embodimentsutilize different ways of improving loop update rate using a twodimensional phase comparator.

The multi-phase comparison of multiple phases derived from a receivedreference signal and multiple phases derived from the local PLL clockmay be generalized into a matrix phase comparator, one embodiment ofwhich is shown in FIG. 11B, with one embodiment of each individual phasecomparator in the matrix shown in FIG. 11A. For descriptive purposes,XOR partial phase comparators arranged in a four by four matrix areillustrated, with no limitation implied by those illustrative choices.Embodiments may be organized into rectangular, square, or sparsematrices of any dimensions M and N, with elements of the matrix beingcomposed of any phase comparator described herein and optionally anyweighting factor computation described herein. As the local clock phasesrepeat cyclically, the leftmost and rightmost columns of the matrixshould be considered as being logically adjacent in their local clockphase relationship. This may be observed in FIG. 11B. Suppose in FIG.11B, CKPLL0=0°, CKPLL1=90°, CKPLL2=180°, and CKPLL3=270°. It should benoted that these numbers are being used purely for illustration. Itwould thus follow that a fifth local oscillator clock CKPLL4 would equal360, which would of course be the same phase as CKPLL0. Thus, theleft-most and right-most columns should be considered adjacent inaccordance with embodiments described above. As used herein, a sparsematrix is any embodiment in which at least one of the described elementweights is zero. In some embodiments, one or more number of referenceclock phases can be compared to one or more than one phases of thefeedback clock. Every extra feedback clock phase (CkPLL_(N)) providesmore detailed information about the phase noise of VCO in time domain.Hence, such a phase comparator may more often provide correction signalsto the loop filter. In other words, a multi-phase feedback systemenables the loop to increase its update rate and correct for phasedeviation of VCO at higher rates. Similarly, if there are more number ofreference clock phases available, higher resolution phase comparison canbe made, and correspondingly correction signals may be applied moreoften in time. If there is only one reference phase, still a controlledchain of delay line (CDL) can be employed to replicate the referenceclock. The bandwidth of control loop for such a delay line needs to beeither very high, or very low in order to guarantee that all the phasesof the replicated clock phases carry similar jitter characteristicswithin the frequency range of interest. A generalized two-dimensionalphase comparator is depicted in FIG. 11B, in which every phase of thefeedback signal can be compared with every phase of the reference clock.

In a full matrix comparison, each of M phases derived from the receivedreference signal is separately phase compared with each of the N phasesderived from the local PLL clock. Each resulting phase error signal isweighted by a configured or predetermined amount, with all (M*N)weighted results summed to produce an aggregate error result. An exampleof one partial phase comparator is shown in FIG. 11A as 1110, composedof XOR phase detector 1112 feeding to result weighting factor 1118. Asshown, each partial phase comparator 1110 receiving CKRef(m) andCkPLL(n) may have a corresponding weighting factor W(m,n) for 0≤m<M−1and 0≤n<N−1. An embodiment of the complete matrix phase comparator 920in FIG. 11B is composed of M*N instances of 1110, each accepting one ofthe M reference phases, herein identified as CkRef0, CkRef1, CkRef2,CkRef3, and one of the N local phase inputs herein identified as CkPLL0,CkPLL1, CkPLL2, CkPLL3, and producing a weighted result e.g. multipleresults 1131, 1132, 1133, 1134 as input to summation 935, which producesan aggregate result 1145.

One familiar with the art will observe that the previously-describedmulti-phase comparator 920 of FIG. 9 is equivalent to apartially-populated instance of the present matrix comparator, i.e.having comparators instantiated across a diagonal of the matrix.Functionally, an identical result may be obtained from a full matrix bysetting the weights along such a diagonal to a nonzero value, and allother comparator weights to zero. It thus follows that other describedbehaviors including simulation of phase offsets, introduction of looptime domain zeroes, etc. may be similarly be obtained by selectiveconfiguration of matrix weighting factors. In such embodiments, eachreference clock phase will be compared with its corresponding feedbackclock phase, i.e. CkRef,m versus CkPLLn. The dynamics of such a diagonalcomparator are similar to a conventional PLL (CPLL) except that theupdate rate is N times faster. Higher loop update rate will allow adiagonal PLL to track input jitter and correct jitter of VCO up tohigher frequencies. In summary, signal (reference) transfercharacteristics, (STF), and VCO noise (jitter) transfer characteristics,(NTF or JTF), of such a system will be N times wider compared with theconventional PLLs.

In at least one embodiment, the matrix comparator of FIG. 11B may besimplified such that different phases of feedback signal (CkPLL) arecompared against only one reference clock phase. Such embodiments areespecially interesting as generally there is only one single referenceclock phase available. In such an implementation W(m,n) are zero unlessm=0.

Assuming a four phase feedback, each phase comparator produces acorrection signal which eventually will be accumulated onto thecapacitor of the loop filter. While under lock condition the integral ofthe correction signal illustrated in the bottom waveform of FIG. 14A iszero, this signal has its main harmonic at 2f_(ref). Any jitter on thereference clock or feedback divider phase will cause some deviations atdifferent edges of the bottom waveform shown in FIG. 14A. The bottomwaveform in FIG. 14A shows the correction signal injected into the loopfilter. Comparing the waveform depicted in the bottom waveform of FIG.14A with the waveform of a simple XOR phase comparator shown in FIG. 15,it is evident that the number of transitions happening at the output ofcomparator during a single reference clock period has been increased bymore than a factor of two. While in both waveforms there are two jumpscreated by the rising and falling edges of CkRef the number oftransitions due to feedback signal has been increased from two in FIG.15 to eight in FIG. 14A (also shown in FIG. 16). Based on this, thefeedback transfer function in a multi-phase diagonal PD will bedifferent form a conventional PD. Due to more number of samples providedby the feedback path in this case, the noise of VCO can be correctedmore often, and hence over a wider frequency bandwidth.

A multi-phase or array phase comparator as described above opens newdoors to design low noise and wide-band PLLs. Described below aredifferent examples of improving performance of PLLs using array phasecomparators.

In some embodiments, array phase comparators provide double edge phasecomparison: Using double edges (rising and falling edges) of thereference clock provides the possibility to make two separate phasecorrections at every cycle. Hence, the BW of PLL can be potentiallyincreased by a factor of two. In case the duty-cycle of the inputreference is not 50%, it will create some ripples at 2·Fref and canincrease deterministic jitter (DJ) of the oscillator. Indeed, as theripple frequency is fairly high, with proper loop filter design it willbe possible to reject major part of this DJ.

In some embodiments, array phase comparators provide Inter-VCO-phasecomparison. To detect and correct duty-cycle and quadrature mismatcherrors (QME), generally designers make comparison between differentphases of a VCO. For such correction, the error signal resulted frominter-VCO-phase comparison is heavily filtered and a very low frequencycorrection signal is applied in a proper point of the system in order tocorrect for duty-cycle distortion or quadrature mismatch. Embodimentsdescribed above may be expanded to detect and correct random jitter ofVCO.

Some embodiments of the two-dimensional discrete-time phase comparatorprovide the possibility to implement a discrete-time filter in front ofPLL. This opportunity can be used to implement special transferfunctions in order to improve the performance of system. For example, byproper choice of digital filter coefficients (e.g., the weightsdescribed above), one may detect and suppress phase noise at specialfrequencies in order to improve tolerance of system against supply orsubstrate noise. Some embodiments provide Phase Interpolation: There aredifferent approaches for rotating phase of oscillator in a PLL andadjusting the exact timing of recovered clock based on systemrequirements. In some embodiments, a diagonal comparator array structureprovides the possibility to rotate a phase of the oscillator withrespect to the phase of reference clock.

Suppose for a given reference clock CkRefm and a given local oscillatorphase CkPLLn, W(m,n)=a for (m−n)=0, W(m,n)=b for |m−n|=1, and zero forthe rest of m and n values (noting that n must be considered modulo thenumber of local oscillator phases, because of its previously-describedcyclic nature. Assuming that a+b=c, and c has a fixed value. In thiscase, by changing a, and b=c−a, it is possible to rotate the phase ofVCO. If lock points corresponding to [a,b]=[c,0] and [a,b]=[0,c] are Taand Tb, respectively, then as depicted in FIG. 17, by changing a (andinversely changing b=c−a) the lock point of oscillator can be adjustedbetween Ta and Tb.

The same argument is valid for any other combination of the twoconsecutive sets of |m−n|=k and |m−n|=k+1, where k is an integer numbersmaller than the size of phase comparator matrices. Suppose a and b arethe weights for two consecutive sets of diagonal comparators k and k+1.If a and b are controlled digitally with two independent sets of N_(b)bits, then the resulted phase interpolator can exhibit N_(b)+1 bits ofresolution. Illustrated in FIG. 18, it can be observed that by properjogging between n(a) and n(b) (control bits corresponding to a and b,respectively), it is possible to add one extra phase point between everytwo phase steps of the original phase interpolator.

An example of diagonal interpolation is given below, where the maindiagonal has a weight a assigned to each element and where an adjacentdiagonal has a weight b assigned to each element:

$\quad\begin{bmatrix}a & b & 0 & 0 \\0 & a & b & 0 \\0 & 0 & a & b \\b & 0 & 0 & a\end{bmatrix}$

Note that due to the cyclical nature of adjacent columns, a weight onthe element of b is included at the bottom-left most element of thematrix. Thus, it always remains that an equal number of elements willcontain a weight of a and b, providing a linear and symmetric phaserelationship, as shown in FIG. 17. Another example of two adjacentdiagonals is given below showing the cyclic nature:

$\begin{matrix}\begin{bmatrix}0 & a & b & 0 \\0 & 0 & a & b \\b & 0 & 0 & a \\a & b & 0 & 0\end{bmatrix} & \;\end{matrix}$

In an alternative embodiment, interpolation can be performed between twoadjacent rows or two adjacent columns of a matrix. The concept is verysimilar to the diagonal embodiment above, in that a first column mayhave a first fixed phase offset, while a second column has a secondfixed phase offset. A weight a may be assigned to each element in thefirst column and a weight b may be assigned to each element in thesecond column, where a +b=c, as described above. Also, it should benoted that the left-most columns and right-most columns should beconsidered adjacent due to the cyclical nature of the local oscillatorclocks. Thus, interpolating the weights a and b in each column willprovide an intermediate phase in between the first and second fixedphases.

Phase Comparator Architecture

The flexibility and architectural simplicity of a matrix comparator maybe offset by the potential complexity of its embodiment, in both thenumber of comparators used to fill an array of M×N elements, and in theimplementation of the required weighting or scaling factor for each ofthose elements.

As a non-limiting example, a PLL utilizing a matrix comparison of eachof eight VCO clock phases against a single reference clock inputincludes eight comparator elements, each associated with a configurableor adjustable scaling factor, and one summation node to produce acombined error result. In one such embodiment shown in FIG. 19A, eachpartial phase comparator 1110 incorporates a multiplying DAC structureto implement the weighting factor as in 1118. In some embodiments, theweighting signal selectively enables one or more switching elements,which may include a transistor switch and a current source, for example.Thus the more switching elements are enabled, a higher weight will beapplied to the partial phase-error signal.

As the distributed capacitance of the DAC structure in the signal pathmay unreasonably degrade signal integrity, an alternative embodimentmoves the DAC out of the signal path. In this second embodiment, a DACelement 1116 is used to scale or adjust supply voltages to a digitalbuffer element 1115 as in FIG. 19B, resulting in a scaled or weightedsignal output.

Circuit applications requiring low power operation may be incompatiblewith resistive DAC usage, thus a further embodiment utilizes switchedcapacitor technology. In this third embodiment of FIG. 19C, the scaledor adjusted supply voltages for buffer 1115 are produced dynamically1117, by dumping measured amounts of charge from one or more sourcecapacitances C1 into the buffering device's power rail capacitance C2.In this illustrative example switch SW1 is shown performing this chargetransfer; switching transistors would be used in a practical embodiment,as well understood in the art, and either or both of C1 and C2 may becomposed of discrete as well as parasitic or distributed capacitances.

An alternative fourth embodiment shown as FIG. 19D does not attempt toadjust or modify the signal output of a single phase comparator, butinstead utilizes a parallel set of phase comparators 1113 whichcollectively produce an aggregate output signal. In such an embodiment,the output drive capability of an individual phase comparator 1113 isconstrained, e.g. by the transistor dimensions of its output driver. Asone example, an aggregate output may be produced by passive summation ofthe individual phase comparator current outputs, and the overall outputamplitude weighted or controlled by enabling or disabling phasecomparators within the set, either individually (as one example,controlled by a thermometer code,) or in groups (as another example,controlled by a binary weighted code.)

During high-speed simulation of that fourth embodiment, transient outputvariations were observed within the gate propagation time of the XORphase detector element, that is, at a finer granularity than the overallbehavior of the XOR gate as a whole. It was noted that an exclusive-ORfunction may be decomposed into distinct NOT-AND-OR logic elements asdescribed by the well-known Boolean equation (x·y)+(x·y), and theobserved behavior suggested that variations among the different currentpaths within the gate implementation were producing the observedvariations. This lead to the embodiment of FIG. 20, where the foursignal paths 2010, 2020, 2030, 2040 are composed of series transistorsforming transistor branches configured to respectively compute thelogical cases x·y, x·y, x·y, x·y, each path further includes anadjustable or configurable impedance which may be a resistor, a currentsource/sink, or in some embodiments may be implemented as a scaling oftransistor geometry to constrain current flow in that signal path. Atiming diagram illustrating the output Iout resulting from inputs X andY is shown in FIG. 21.

Adjustment of the four signal path impedances by introducing pathresistance, scaling transistor dimensions, or directly adjusting currentflows may be used to control output amplitude within the overallresponse of the XOR gate as a whole, thus producing the desired weighingfunction. As one example, consider each signal path impedance in FIG. 20as being composed of four parallel resistors each controlled by atransistor switch. In such a case, a portion of a weighting signal whichmay be a four-bit thermometer code t₀-t₃ (as one example offered withoutlimitation) can enable four distinct levels of current flow withinsignal path 1920, which may be seen in FIG. 21 to adjust one segment2110 of the overall output Iout. Continuing this example, t₄-t₇ adjustssegment 2120, t₈-t₁₁ adjusts segment 2130, and t₁₂-t₁₅ adjusts segment2140. Although in some embodiments the weights of each of the foursegments will be adjusted identically, this is not required. Asexamples, t₀-t₃ and t₄-t₇ may be configured to different values thant₈-t₁₁ and t₁₂-t₁₅ to provide increased overall adjustment granularityas previously described relative to FIG. 5. As another example, t₀-t₃and t₈-t₁₁ may be configured to different values than t₄-t₇ and t₁₂-t₁₅to intentionally introduce a DC offset at the output.

Independently adjusting the four segments of each XOR operation may alsofacilitate the previously-described matrix comparator operations,including interpolation. As one example, two XOR comparators as in FIG.20 with outputs connected to a common summation node may be used tocompare a reference clock with two local clock phases, as previouslydescribed. If interpolation control values a, b, c, d represent theweighting signals applied to the first XOR segments (t₀-t₃, t₄-t₇,t₈-t₁₁, t₁₂-t₁₃), and 1-a, 1-b, 1-c, 1-d represent the weighting signalsapplied to the second XOR segments, the common summation node resultcorresponds to an interpolation between the first and second local clockphase comparisons, with the interpolation control values permittingadjustment of the effective clock phase.

Alternatively, scaled transistors or explicit current source/sinkcircuits may be used instead of resistors to control current flow; as inthe resistive example, different numbers of enabled and disabledparallel current paths may be used to adjust the overall output Iout.the portions of the weighting signal above collectively form theweighting signal t₀-t₁₅ for the corresponding partial phase comparator.

The adjustable or configurable elements described herein may be combinedwith or equivalently be substituted by comparable known art elements,including without limitation R-2R ladder structures controlled bytransistor switches, resistive chain structures controlled by transistorswitches, equal-value or binary-weighted resistors configured in series-or parallel-connected combination by transistor switches, and fixedand/or configurable current sources and current sinks.

Specific values and quantities described in examples are provided tofacilitate explanation, without implying limitation.

Simulations

The steady state response of two different types of phase comparatorsare shown in FIG. 22. In these simulations XOR based phase comparatorcells have been utilized. The gain of a diagonal phase comparator versusinput phase difference is very similar to a simple XOR phase comparator.The main difference between the two phase comparators is that errorsignal produced by a diagonal phase comparator is distributed in time.The row phase comparator, however, exhibits a different response. As canbe seen, depending on which row of the matrix comparator has beenselected, the response exhibits a different shift in time. FIG. 22 showsthe response for two different cases.

An MCPLL (matrix phase comparator based CPLL) has been designed in aconventional 28-nm CMOS technology. A current steering architecture withK_(VCO)≈20 GHz/V is selected for the VCO. A loop filter as shown in FIG.24 is chosen to be R1=100Ω in series to C1=20 pF with ICPC=100 μA. Thereference clock frequency is 6.25 GHz with N_(div)=1. The XOR CPLL isbased on 2R×2F architecture (comparing 2 reference phases with 2feedback phases at every reference clock cycle). For the implementedMCPLL, a 2R×4F architecture has been chosen.

FIG. 23 shows the transfer characteristics of the two PLLs simulated intransistor level. As shown, the input signal is sampled at 2fref, hencethe system Nyquist rate is fref. Because of that, as depicted in FIG.22, both PLLs have a null at fref=6.25 GHz. The parasitic pole of theVCO in addition to the parasitic pole at the output of CPC cause 1.2 and2.4 dB peaking in CPLL and MPLL, respectively, for the two PLLs. It canbe seen also that the MCPLL transfer characteristic stays closer to 0 dBbetween 100 MHz to 1.2 GHz. The cutoff frequency for the two PLLs are2.48 (=40% of fref) and 5.02 GHz (=80% of fref), respectively.Therefore, it can provide better jitter tracking. Moreover, MCPLLgenerates 55 fs of rms jitter, while jitter generation for CPLL is 79 fsrms. Due to higher BW, the phase noise of VCO in MCPLL is filtered outover a wider range and because of that this architecture exhibitsconsiderably lower jitter generation, comparable to performance of LVVCO based PLLs.

Data-Driven Phase Comparator

In some embodiments, the above-described matrix phase comparatorarchitecture may be applied to extraction of a clock signal fromtransitions occurring on multiple data sub-channels, as one example onthe various sub-channels of a vector signaling code communicated over amulti-wire bus. In such embodiments, vector signaling codes withguaranteed transition density over time resulting from bit-levelencoding, or provided by vector encoding schemes such as taught by[Shokrollahi I] are amenable to such combination. FIG. 25 illustrates areceiver utilizing a data-driven phase comparator, in which data signalsreceived in parallel from MIC0-MIC4 are monitored for transitions,providing clock phase information to Clock Recovery 2600 and thuscontrolling the phase of sampling clocks ph000, ph090, ph180, ph270.FIG. 26 provides a more detailed view of Clock Recovery 2600, whichutilizes matrix phase comparator 2610.

Considering the multiple data inputs being monitored by such a system,several operational considerations are apparent. First, as any receiveddata bit may remain in either the “1” or “0” state in consecutive unitintervals, only data transitions between those states are relevant toPLL phase. In particular, between any two consecutive unit intervals atransition may or may not occur in any given data bit; indeed, notransition may occur on any data bit in a given clock interval. If atransition does occur, the matrix phase comparator may utilize thetiming of that transition to update the PLL clock phase, while if notransition occurs, the PLL clock may be allowed to continue unchanged.If two or more data lines transition in the same clock interval, thetiming errors derived from each such transition may be summed, which isconsistent with the previously-described matrix phase comparatorbehavior.

This behavior suggests that a state-machine phase detector may be asuitable candidate for the phase comparison elements of the comparisonmatrix, as such designs may be configured to respond only to signaltransitions rather than signal level, and may be configured to output a“no change” result in the absence of a signal transition. In someembodiments, the partial phase comparators 2712 may take the form ofedge-triggered bang-bang detectors configured to generate partialphase-error signals in response to determining a transition occurred. Ablock diagram of an exemplary edge-triggered bang-bang phase detector isshown in FIG. 29. In some embodiments, the partial phase comparators maytake the form of linear edge-triggered phase detectors to generatepartial phase-error signals in response to determining a transitionoccurred. A block diagram of a linear edge-triggered phase detector isillustrated in FIG. 30, and a corresponding waveform is shown in FIG.31.

Another embodiment may incorporate a data signal transition detectors,one example including an XOR gate comparing a data signal with aslightly time delayed copy of the same data signal, for example afterpassing through a logic buffer gate. Such an embodiment will output alogical pulse at each transition, and the edge of such pulse may bephase compared to a PLL clock edge, using any phase detector aspreviously described. An advanced embodiment may further incorporate agating or time windowing function in the partial phase detectors toproduce a “no change” error result from any partial phase detector notreceiving a data signal transition in a given time interval.

FIG. 26 illustrates one embodiment of a multi-line Clock Recovery 2600.MIC0-MIC4 are the detected vector signaling code sub-channels (i.e.decoded data bits) for a code similar to that used in the example ofFIG. 2. Any transition on a detected vector signaling code sub-channelmay produce a partial phase-error signal relative to one or more localclock phases, each such partial phase-error signal being (in thisexample) an analog signal either pulsing up, pulsing down, or remainingunchanged. The summation 2650 of all partial phase-error signals is lowpass filtered 2660, with the result used to adjust the VCO 2670frequency. In some embodiments, if the transmission medium hassignificantly different propagation velocity for different propagationmodes, various sub-channels may experience eye-closures, resulting ininter-subchannel skew. In order to offset such inter-subchannel skew,the phase interpolator 2690 may be configured to independently adjustphases of each local oscillator signal according to the associated datasignal in order to correct for such sub-channel specific skew.Alternatively, analog delay elements (not shown) may be used tointroduce sub-channel specific delay to compensate for sub-channelspecific skew.

FIG. 27 further illustrates one embodiment of Matrix Phase Detector2610, as including an array of component phase detectors 2710. As oneexample not implying limitation, each partial phase detector 2710 may inturn include an edge-sensitive state machine phase detector, chargepump, and configurable weighting function, as described in [Tajalli IV].Matrix phase detector 2610 thus compares each of detected data signalsMIC0-MIC4 to four phases of local PLL clock ph0, ph090, ph180, ph270,with each such comparison 2712 producing a phase comparison result thatis subsequently weighted 2715. In one particular embodiment, theweighted results are produced as analog currents, thus all such results2720, 2730, 2740, 2750, 2760 may be summed by mutual connection at acurrent summation node 2650, producing aggregate result 2655 directly.As will be clear to one familiar with the art, comparable results mayalso be produced by explicit summation of voltages, numericalcomputation, etc. thus no limitation is implied. In some embodiments,partial phase comparator 2710 may further receive the Transition_ENenable signal (not shown) to selectively output the correspondingpartial phase error signal. In some embodiments, each partial phasecomparator 2710 receives a corresponding Transition_EN(m) signalassociated with the corresponding detected data signal received at thepartial phase comparator 2710. For example, MIC0 may have an associatedenable signal Transition_EN0, MIC1 may have an associated enable signalTransition_EN1, etc.

The weights of the individual matrix comparison elements 2710 of matrixphase comparator 2610 may be set uniformly (i.e. with a transition onany data signal equally affecting all clock phases,) or non-uniformlysuch that particular clock phases are more or less affected. Asdescribed in [Tajalli IV], other effects including simulation of phaseoffsets, introduction of loop time domain zeroes, etc. may be obtainedby selective configuration of matrix weighting factors.

FIG. 26 shows one embodiment of multi-line Clock Recovery 2600.MIC0-MIC4 are the detected vector signaling code sub-channels (i.e.detected data signals or decoded data bits after slicing and sampling)for a code similar to that used in the example of FIG. 2. Any transitionon a detected data signal may produce a phase error relative to one ormore local clock phases, each such phase error being (in this example)an analog signal either pulsing up, pulsing down, or remainingunchanged. The summation 2650 of all phase errors is low pass filtered2660, with the result used to adjust the VCO 2670 frequency.

FIG. 28 is a flowchart of a method 2800, in accordance with someembodiments. As shown, method 2800 includes receiving 2802, at adata-driven phase comparator circuit, a plurality of data signals inparallel from a plurality of multi-input comparators (MICs) connected toa multi-wire bus, wherein at least one MIC is connected to at leastthree wires of the multi-wire bus, and one or more phases of a localoscillator signal, the data-driven phase comparator circuit comprising aplurality of partial phase comparators, generating 2804, a plurality ofpartial phase-error signals using the partial phase comparators, eachpartial phase-error signal generated by receiving (i) a correspondingphase of the local oscillator signal and (ii) a corresponding datasignal of the plurality of data signals and responsive to adetermination that a transition occurred in the corresponding datasignal, generating the partial phase-error signal based on a comparisonof the corresponding phase of the local oscillator signal and thecorresponding data signal, and generating 2806 a composite phase-errorsignal by summing the plurality of partial phase error signals, thecomposite phase-error signal for setting a local oscillator generatingthe one or more phases of the local oscillator signal in a lockcondition.

In some embodiments, the partial phase-error signals are analog signalsformed using respective charge pump circuits. In such embodiments, themethod further includes filtering the composite phase-error signal.

In some embodiments, the method further includes introducing, for agiven partial phase comparator, a sub-channel specific delay into thecorresponding phase of the local oscillator signal, the sub-channelspecific delay associated with the data signal received at the givenpartial phase comparator.

In some embodiments, the comparison of the corresponding phase of thelocal oscillator signal and the corresponding data signal is formedusing a linear edge-triggered phase detector. Alternatively, thecomparison of the corresponding phase of the local oscillator signal andthe corresponding data signal may be formed using an edge-triggeredbang-bang phase detector.

In some embodiments, the method further includes applying a weight tothe partial phase-error signal. In some embodiments, the plurality ofdata signals have a collective transition density above a predeterminedthreshold. In some embodiments, the method further includes outputting ano-change result in response to determining no transition occurred. Insuch embodiments, outputting the no-change result includes setting thepartial phase comparator in a high-impedance state.

PLL Startup

In many communications environments, the minimum transition density ofdata signals may be low, thus for relatively long periods of time thePLL oscillator must continue running with no change in frequency. Aspreviously described, selection of a phase detector design capable ofbeing configured to produce a “no change” output in the absence of datatransitions helps satisfy this requirement, as does selection of anoscillator with good stability characteristics.

At startup, however, the situation may be quite different. The VCO maystart oscillation at very high or very low frequency relative to itsdesired operating frequency, and the density of transitions received ondata lines may be very high, especially if an initialization proceduretransmits training patterns or other special data sequence as part ofCTLE adjustment or other receiver calibration. Thus, it is possible thatPLL lock may take considerable time, or may result in VCO operation atthe wrong frequency.

[Tajalli III] describes a “Frequency Lock Assist” for PLLinitialization, consisting of an additional phase/frequency detectorthat can override spurious frequency excursions and force the PLL into anormal operational mode, at which point its normal phase comparator maytake over.

Alternative Embodiments

The clock signal received from MIC5 in FIG. 2 after being transportedover two dedicated clock wires could just as easily be received from, asone example MIC4, having been transported as one sub-channel of thevector signaling code also carrying the data. This method of embeddingthe clock in a vector signaling code sub-channel is described in[Shokrollahi II] and [Holden III]. All of the described clock embeddingembodiments therein may be beneficially combined with the PLL and timingcontrol mechanisms described herein, without limitation.

We claim:
 1. An apparatus comprising: a plurality of multi-inputcomparators (MICs) configured to generate a set of MIC outputs byforming respective orthogonal linear combinations of a set of wiresignals received in parallel over a plurality of wires of a multi-wirebus, at least one MIC output having been pre-coded with a transitiondensity code and at least one MIC connected to at least three wires; adata-driven phase comparator circuit configured to receive the pluralityof MIC outputs in parallel from the plurality of MICs and one or morephases of a local oscillator signal, the data-driven phase comparatorcircuit comprising a plurality of partial phase comparators, eachpartial phase comparator configured to: receive (i) a correspondingphase of the local oscillator signal and (ii) a corresponding MIC outputof the plurality of MIC outputs; responsive to a determination that atransition occurred in the corresponding MIC output, generate a partialphase-error signal based on a comparison of the corresponding phase ofthe local oscillator signal and the corresponding MIC output; apply arespective weighting factor to the partial phase-error signal, whereinat least two partial phase comparators of the plurality of partial phasecomparators have different weighting factors; and a summation circuitconfigured to receive each partial phase-error signal and to generate acomposite phase-error signal to set a local oscillator generating theone or more phases of the local oscillator signal in a lock condition.2. The apparatus of claim 1, wherein each partial phase comparatorcomprises a charge pump configured to generate the partial phase-errorsignal in analog form.
 3. The apparatus of claim 2, further comprising aloop filter connected to the plurality of charge-pump circuits, the loopfilter configured to receive the composite phase-error signalrepresenting an analog summation of the plurality of partial phase-errorsignals, and to output a filtered phase-error signal.
 4. The apparatusof claim 1, wherein each partial phase comparator further comprises adelay element, the delay element configured to introduce a sub-channelspecific delay into the corresponding phase of the local oscillatorsignal.
 5. The apparatus of claim 1, wherein each partial-phasecomparator comprises a linear edge-triggered phase detector configuredto form the comparison of the corresponding MIC output and thecorresponding phase of the local oscillator signal.
 6. The apparatus ofclaim 1, wherein each partial-phase comparator comprises anedge-triggered bang-bang phase detector configured to form thecomparison of the corresponding MIC output and the corresponding phaseof the local oscillator signal.
 7. The apparatus of claim 1, wherein thetransition density code is an 8b10b code.
 8. The apparatus of claim 1,wherein the transition density code is a 64b66b code
 9. The apparatus ofclaim 1, wherein the partial phase-error signal generated from the MICoutput having been pre-coded with the transition density code has alarger weighting factor than at least one other partial phase-errorsignal generated from another MIC output.
 10. The apparatus of claim 1,wherein each partial phase-error signal comprises a plurality ofweighted segments, each partial-phase comparator comprises a pluralityof logic branches, each logic branch configured to generate a respectiveweighted segment of the plurality of weighted segments according to arespective input logic combination between the corresponding MIC outputand the corresponding phase of the local oscillator signal, each logicbranch having an independently configurable weighting factor.
 11. Amethod comprising: generating a set of MIC outputs using a plurality ofmulti-input comparators (MICs), each MIC output generated by forming arespective orthogonal linear combination of a set of wire signalsreceived in parallel over a plurality of wires of a multi-wire bus, atleast one MIC output having been pre-coded with a transition densitycode and at least one MIC connected to at least three wires; generatinga plurality of partial phase-error signals using a plurality of partialphase comparators, wherein generating each partial phase-error signalcomprises: receiving (i) a corresponding phase of the local oscillatorsignal and (ii) a corresponding MIC output of the plurality of MICoutputs; responsive to a determination that a transition occurred in thecorresponding MIC output, generating a comparison of the correspondingphase of the local oscillator signal and the corresponding MIC output;generating the partial phase-error signal by applying a respectiveweighting factor to the comparison of the corresponding phase of thelocal oscillator signal and the corresponding MIC output, wherein atleast two partial phase-error signals have different weighting factors;and generating a composite phase-error signal using a summation circuitby adding the plurality of partial phase-error signals, the compositephase-error signal setting a local oscillator generating the one or morephases of the local oscillator signal in a lock condition.
 12. Themethod of claim 11, wherein each partial phase-error signal is generatedas an analog signal at an output of a charge pump, and wherein thecomposite phase-error signal is generated as an analog summation of theplurality of partial phase-error signals.
 13. The method of claim 11,further comprising filtering the composite phase-error signal.
 14. Themethod of claim 11, wherein generating each partial phase-error signalfurther comprises applying a sub-channel specific delay to thecorresponding phase of the local oscillator signal.
 15. The method ofclaim 11, wherein each partial phase-error signal is generated using alinear edge-triggered phase detector by forming the comparison of thecorresponding MIC output and the corresponding phase of the localoscillator signal.
 16. The method of claim 11, wherein eachpartial-phase comparator comprises an edge-triggered bang-bang phasedetector configured to form the comparison of the corresponding MICoutput and the corresponding phase of the local oscillator signal. 17.The method of claim 11, wherein the transition density code is an 8b10bcode.
 18. The method of claim 11, wherein the transition density code isa 64b66b code
 19. The method of claim 11, wherein the partialphase-error signal generated from the MIC output having been pre-codedwith the transition density code has a larger weighting factor than atleast one other partial phase-error signal generated from another MICoutput.
 20. The method of claim 11, wherein each partial phase-errorsignal comprises a plurality of weighted segments, each partial-phasecomparator comprises a plurality of logic branches, each logic branchgenerating a respective weighted segment of the plurality of weightedsegments according to a respective input logic combination between thecorresponding MIC output and the corresponding phase of the localoscillator signal, each logic branch further applying an independentlyconfigurable weighting factor to the weighted segment.